HIP6014

FN4420 Rev.2.00

Page 9 of 12

March 2000

small case size perform better than a single large case

capacitor.

Output Inductor Selection

The output inductor is selected to meet the output voltage

ripple requirements and minimize the converter’s response

time to the load transient. The inductor value determines the

converter’s ripple current and the ripple voltage is a function of

the ripple current. The ripple voltage and current are

approximated by the following equations:

Increasing the value of inductance reduces the ripple current

and voltage. However, the large inductance values reduce the

converter’s response time to a load transient.

One of the parameters limiting the converter’s response to a

load transient is the time required to change the inductor current.

Given a sufficiently fast control loop design, the HIP6014 will

provide either 0% or 100% duty cycle in response to a load

transient. The response time is the time required to slew the

inductor current from an initial current value to the transient

current level. During this interval the difference between the

inductor current and the transient current level must be supplied

by the output capacitor. Minimizing the response time can

minimize the output capacitance required.

The response time to a transient is different for the application

of load and the removal of load. The following equations give

the approximate response time interval for application and

removal of a transient load:

where: ITRAN is the transient load current step, tRISE is the

response time to the application of load, and tFALL is the

response time to the removal of load. With a +5V input source,

the worst case response time can be either at the application

or removal of load and dependent upon the DACOUT setting.

Be sure to check both of these equations at the minimum and

maximum output levels for the worst case response time. With

a +12V input, and output voltage level equal to DACOUT, tFALL

is the longest response time.

Input Capacitor Selection

Use a mix of input bypass capacitors to control the voltage

overshoot across the MOSFETs. Use small ceramic

capacitors for high frequency decoupling and bulk capacitors

to supply the current needed each time Q1 turns on. Place

the small ceramic capacitors physically close to the

MOSFETs and between the drain of Q1 and the source of Q2.

The important parameters for the bulk input capacitor are the

voltage rating and the RMS current rating. For reliable

operation, select the bulk capacitor with voltage and current

ratings above the maximum input voltage and largest RMS

current required by the circuit. The capacitor voltage rating

should be at least 1.25 times greater than the maximum input

voltage and a voltage rating of 1.5 times is a conservative

guideline. The RMS current rating requirement for the input

capacitor of a buck regulator is approximately 1/2 the DC load

current.

For a through hole design, several electrolytic capacitors

(Panasonic HFQ series or Nichicon PL series or Sanyo MV-GX

or equivalent) may be needed. For surface mount designs,

solid tantalum capacitors can be used, but caution must be

exercised with regard to the capacitor surge current rating.

These capacitors must be capable of handling the surge-

current at power-up. The TPS series available from AVX, and

the 593D series from Sprague are both surge current tested.

MOSFET Selection/Considerations

The HIP6014 requires 2 N-Channel power MOSFETs. These

should be selected based upon rDS(ON), gate supply

requirements, and thermal management requirements.

In high-current applications, the MOSFET power dissipation,

package selection and heatsink are the dominant design

factors. The power dissipation includes two loss components;

conduction loss and switching loss. The conduction losses are

the largest component of power dissipation for both the upper

and the lower MOSFETs. These losses are distributed

between the two MOSFETs according to duty factor (see the

equations below). Only the upper MOSFET has switching

losses, since the Schottky rectifier clamps the switching node

before the synchronous rectifier turns on. These equations

assume linear voltage-current transitions and do not

adequately model power loss due the reverse-recovery of the

lower MOSFET’s body diode. The gate-charge losses are

dissipated by the HIP6014 and don't heat the MOSFETs.

However, large gate-charge increases the switching interval,

tSW which increases the upper MOSFET switching losses.

Ensure that both MOSFETs are within their maximum junction

temperature at high ambient temperature by calculating the

temperature rise according to package thermal-resistance

specifications. A separate heatsink may be necessary

depending upon MOSFET power, package type, ambient

temperature and air flow.

Standard-gate MOSFETs are normally recommended for use

with the HIP6014. However, logic-level gate MOSFETs can be

used under special circumstances. The input voltage, upper

gate drive level, and the MOSFET’s absolute gate-to-source

voltage rating determine whether logic-level MOSFETs are

appropriate.

I=

VIN - VOUT

Fs x L

VOUT

VIN

VOUT = I x ESR



tRISE =

L x ITRAN

VIN - VOUT

tFALL =

L x ITRAN

VOUT

PUPPER = Io2 x rDS(ON) x D +

1

2

Io x VIN x tSW x FS

PLOWER = Io2 x rDS(ON) x (1 - D)

Where: D is the duty cycle = VOUT / VIN,

tSW is the switch ON time, and

FS is the switching frequency.