ISL6334DIRZ datasheet(23/28 Pages) RENESAS

23 / 28 page

ISL6334D

FN6802 Rev 4.00

Page 23 of 28

Apr 29, 2016

schematics, bills of materials and example board layouts for all

common microprocessor applications.

Power Stages

The first step in designing a multiphase converter is to determine

the number of phases. This determination depends heavily upon

the cost analysis, which in turn depends on system constraints

that differ from one design to the next. Principally, the designer

will be concerned with whether components can be mounted on

both sides of the circuit board; whether through-hole components

are permitted; and the total board space available for power

supply circuitry. Generally speaking, the most economical

solutions are those in which each phase handles between 15A

and 25A. All surface mount designs will tend toward the lower

end of this current range. If through-hole MOSFETs and inductors

can be used, higher per-phase currents are possible. In cases

where board space is the limiting constraint, current can be

pushed as high as 40A per phase, but these designs require heat

sinks and forced air to cool the MOSFETs, inductors and

heat-dissipating surfaces.

MOSFETs

The choice of MOSFETs depends on the current each MOSFET will

be required to conduct; the switching frequency; the capability of

the MOSFETs to dissipate heat; and the availability and nature of

heatsinking and air flow.

LOWER MOSFET POWER CALCULATION

The calculation for heat dissipated in the lower MOSFET is

simple, since virtually all of the heat loss in the lower MOSFET is

due to current conducted through the channel resistance

(rDS(ON)). In Equation 21, IM is the maximum continuous output

current; IP-P is the peak-to-peak inductor current (see

Equation 1); d is the duty cycle (VOUT/VIN); and L is the

per-channel inductance.

An additional term can be added to the lower-MOSFET loss

equation to account for additional loss accrued during the dead

time when inductor current is flowing through the lower-MOSFET

body diode. This term is dependent on the diode forward voltage

at IM, VD(ON); the switching frequency, fsw; and the length of

dead times, td1 and td2, at the beginning and the end of the

lower-MOSFET conduction interval respectively.

Thus the total maximum power dissipated in each lower MOSFET

is approximated by the summation of PLOW,1 and PLOW,2.

UPPER MOSFET POWER CALCULATION

In addition to rDS(ON) losses, a large portion of the upper MOSFET

losses are due to currents conducted across the input voltage (VIN)

during switching. Since a substantially higher portion of the upper

MOSFET losses are dependent on switching frequency, the power

calculation is more complex. Upper MOSFET losses can be divided

into separate components involving the upper MOSFET switching

times; the lower MOSFET body-diode reverse-recovery charge, Qrr;

and the upper MOSFET rDS(ON) conduction loss.

When the upper MOSFET turns off, the lower MOSFET does not

conduct any portion of the inductor current until the voltage at

the phase node falls below ground. Once the lower MOSFET

begins conducting, the current in the upper MOSFET falls to zero

as the current in the lower MOSFET ramps up to assume the full

inductor current. In Equation 23, the required time for this

commutation is t1 and the approximated associated power loss

is PUP,1.

At turn-on, the upper MOSFET begins to conduct and this

transition occurs over a time, t2. In Equation 24, the approximate

power loss is PUP,2.

A third component involves the lower MOSFET’s reverse recovery

charge, Qrr. Since the inductor current has fully commutated to

the upper MOSFET before the lower MOSFET’s body diode can

draw all of Qrr, it is conducted through the upper MOSFET across

VIN. The power dissipated as a result is PUP,3 and is

approximated in Equation 25:

Finally, the resistive part of the upper MOSFET’s is given in

Equation 26 as PUP,4.

The total power dissipated by the upper MOSFET at full load can

now be approximated as the summation of the results from

Equations 23, 24, and 25. Since the power equations depend on

MOSFET parameters, choosing the correct MOSFETs can be an

iterative process involving repetitive solutions to the loss

equations for different MOSFETs and different switching

frequencies, as shown in Equation 26.

Current Sensing Resistor

The resistors connected to the ISEN+ pins determine the gain in

the channel-current balance loop and set the overcurrent trip

point. Select values for these resistors by using Equation 27:

Where RISEN is the sense resistor connected to the ISEN+ pin, N

is the active channel number, RX is the resistance of the current

sense element, either the DCR of the inductor or RSENSE

depending on the sensing method, and IOCP is the desired

overcurrent trip point. Typically, IOCP can be chosen to be 1.2x

the maximum load current of the specific application.

With integrated temperature compensation, the sensed current

signal is independent on the operational temperature of the

PLOW 1



rDS ON



------





 2

–



ILP-P



2 1d

–



------------------------------------

(EQ. 21)

PLOW 2



VDON

 fsw

------

IP-P

----------



 t

------

IP-P

----------

–







td2

(EQ. 22)

PUP 1, VIN

------

IP-P

----------



 t1

----









(EQ. 23)

PUP 2

VIN

------

IP-P

----------

–





 t2

----









(EQ. 24)

PUP 3,

VIN Qrr fS

(EQ. 25)

PUP 4, rDS ON



------





 2

IP-P

----------d



(EQ. 26)

RISEN

105 10 6

–



---------------------------

IOCP

--------------

(EQ. 27)

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